Wideband wireless systems are increasingly prevalent in modern communication devices. Applications of wideband wireless systems include high-bandwidth digital communication devices such as 3G and 4G mobile telephony systems, wireless local area network or “Wi-Fi”-enabled systems, and television broadcast systems. The successful operation of such systems often is dependent on accurate radio frequency (RF) filtering.
In a typical wideband wireless system, an antenna is used to receive the entire wireless spectrum, which contains the desired RF signal as well as several undesired blockers. These undesired blockers include adjacent blockers near the frequency of the desired signal and far-out blockers farther away from the frequency of the desired RF signal. High-Q filters are typically tuned to perform bandpass filtering around the desired RF signal. High-Q filters typically are composed of discrete passive devices, such as resistors, capacitors, inductors, and varactors. This configuration mainly serves to attenuate the far-out blockers. The tuning of a high-Q filter typically is performed by means of a feedback signal from a RF tuner component, whereby the accuracy of the filter depends on the accuracy of the feedback signal. Moreover, the sharpness of the filter is a trade-off with the degree to which the desired signal is permitted to droop. The RF tuner component includes a low-noise amplifier (LNA) that amplifies the signal level in such a way that it is sufficiently above the noise floor of the successive blocks of the RF tuner chip. The resulting amplified signal is then fed into a down-conversion mixer that frequency translates the desired RF signal into an intermediate frequency (IF) signal. After down conversion, the IF signal is then further filtered by sharp low frequency filters (at an IF frequency) to attenuate the adjacent blockers.
Although filtering at IF typically can be performed more efficiently than filtering at RF, there are numerous reasons for filtering at RF. For one, RF filtering helps to reduce the total input power to the LNA of the RF tuner component by rejecting far-out blockers, which enhances the effective linearity of the LNA. In many instances, RF surface acoustic wave (SAW) filters or discrete RLC-based filters are used to provide this RF filtering. Such implementations, however, tend to increase overall system costs.
Another reason for RF filtering is image rejection. In a typical RF-to-IF conversion process, a single down-converter mixer down converts the RF spectrum into an IF signal using a local oscillator (LO). The LO typically is a periodic signal having a primary frequency fLO and which typically is generated by an on-chip device, such as a phase-locked loop (PLL). The down-converter mixer produces frequency terms that are the sums and differences between the positive and negative values of the frequencies of signals found in the RF spectrum, including the frequency fCH of the desired RF signal and the frequency fBL of an undesired blocker. The relevant frequency terms are the difference products fCH-fLO, fLO-fCH, fBL-fLO, and fLO-fBL. Assuming that a low-pass filter (LPF) following the down-converter mixer attenuates the sum terms produced by the down-converter mixer and assuming that the separation in frequency between the desired RF signal and the undesired blocker to the LO are equal to one another, the IF spectrum will be composed of the desired signal overlapping in frequency, or smearing, with the undesired blocker. Accordingly, when an undesired blocker satisfies the condition that the separations in frequency between the desired RF signal and between the undesired blocker and the LO are equal to one another, the undesired blocker is said to lie in the image frequency of the desired signal. Image rejection then becomes the process of inhibiting the RF content at the image frequency or canceling the RF energy at the image frequency when down converting to an IF signal.
One conventional approach for image rejection relies on a dual-conversion architecture, or a heterodyne, architecture. In this architecture, two mixers are utilized. The first mixer converts the RF spectrum into an initial IF spectrum. A high frequency for the initial IF spectrum results in a greater separation between the LO frequency and the RF signal, and thus the image also is further in frequency from the LO frequency. This greater separation thus enables a reasonably low-cost filter to be used to filter out the image. Once the image has been removed, the second mixer is then required to frequency translate the resulting signal into the desired final IF spectrum. Such topologies, however, require two high-performance local oscillators, one for each of the two mixers, and careful frequency planning is needed to avoid undesired overlap of mixing terms between the two mixers. Another conventional approach for image rejection implements a complex image reject mixer that cancels the image through the appropriate phase subtractions using a resistive-capacitive (RC) polyphase filter after the down-conversion mixer. Device mismatch of resistors and capacitors limits the performance of this approach.
Another reason for filtering in the RF domain is to filter out the spectrum near the harmonics of the LO signal. This process is commonly referred to as harmonic reject filtering. The need for harmonic reject filtering is particularly acute when the LO signal is a square waveform, which has strong odd order harmonic terms. When driven by a square wave LO, the down-converter mixer downconverts the RF spectrum near the LO frequency to IF, as well as downconverting the spectrum near the odd-order harmonics of the LO frequency to IF. Accordingly, any blockers near the odd-order harmonics of the LO frequency, and particularly the third-order and fifth-order harmonics, will fold onto the desired signal. This is usually addressed by having sufficient RF filtering before the mixer. This situation can be better understood by considering Equation 1 below, which is a Fourier Series representation of a square wave form:
                              sq          ⁡                      (            f            )                          =                              4            π                    ⁢                                    ∑                              n                =                0                            ∞                        ⁢                                          1                                                      2                    ⁢                    n                                    +                  1                                            ⁢                              sin                ⁡                                  (                                                            2                      ⁢                      n                                        +                    1                                    )                                            ⁢              2              ⁢              π              ⁢                                                          ⁢                              f                0                            ⁢              t                                                          EQ        .                                  ⁢        1            where t is time (in seconds), f0 is the LO frequency (also in Hz). As Equation 1 illustrates, a square wave can be represented as an infinite sum of sinusoidal signals operating at odd order harmonics of the LO frequency scaled by progressively decreasing coefficients. This demonstrates that blockers near the third-order harmonic (that is, 3*fLO) and the fifth-order harmonic (that is, 5*fLO) would be downconverted by the mixer to IF and would add to the desired signal at IF, effectively smearing the desired signal.